A power supply used in cooking appliances based on high-frequency heating such as a microwave oven used at home has been required to be small in size and light in weight owing to the nature of the cooking appliances. It is desirable that the space for accommodating the power supply is small in order to easily carry it and enlarge a cooking space in the kitchen. For this reason, the microwave oven is becoming smaller and lighter and being manufactured at low cost with employing a switching power supply. As a result, the power supply outputs a current waveform containing lots of harmonic components which are generated by a switching operation of the power supply. In addition, the microwave oven consumes as much as 2000 watts for the sake of shortening the cooking time. As a result, an absolute value of the current is also increased, and it makes difficult to meet a harmonics performance of the power supply. In light of this problem, a control method (improvement measure) for preventing generation of the harmonic current components has been proposed (for example, see Patent Document 1).
FIG. 9 shows one exemplary diagram of a magnetron-driving power supply for a high frequency heating apparatus (inverter power supply). The magnetron-driving power supply is constituted by a direct-current (DC) power supply 1, a leakage transformer 2, a first semiconductor switching element 3, a first capacitor 5 (snubber capacitor), a second capacitor 6 (resonant capacitor), a third capacitor 7 (smoothing capacitor) a second semiconductor switching element 4, a driving unit 13, a full-wave voltage doubler rectification circuit 11, and a magnetron 12.
The DC power supply 1 applies a DC voltage VDC to a serially connected circuit including the second capacitor 6 and a first coil winding 8 of the leakage transformer 2 by performing a full-wave rectification of a commercial power supply. The first semiconductor switching element 3 and the second semiconductor switching element 4 are connected to each other in series and the serially connected circuit including the second capacitor 6 and the first coil winding 8 of the leakage transformer 2 is connected in parallel to the second semiconductor switching element 4.
The first capacitor 5 is connected in parallel to the second semiconductor switching element 4 and serves as the snubber that prevents a surging current (voltage) during a switching process. The high AC voltage output generated in a second coil winding 9 of the leakage transformer 2 is transformed into a high DC voltage in the full-wave voltage doubler rectification circuit 11, and then applied between the anode and cathode of the magnetron 12. A third coil winding 10 of the leakage transformer 2 supplies current to the cathode of the magnetron 12.
The first semiconductor switching element 3 and the second semiconductor switching element 4 are each constituted by an IGBT and a flywheel diode connected in parallel to the IGBT. As a matter of course, the first and second semiconductor switching elements 3 and 4 are not limited to such a kind, but a thyristor, a GTP switching device, and the like can be also used.
The driving unit 13 has an oscillation unit therein for generating driving signals for driving the first semiconductor switching element 3 and the second semiconductor switching element 4. The oscillation unit generates a square wave with a predetermined frequency and transmits the driving signals to the first semiconductor switching element 3 and the second semiconductor switching element 4. Immediately after any one of the first semiconductor switching element 3 and the second semiconductor switching element 4 is turned off, voltage across the both ends of the other semiconductor switching element is high. Consequently, when any one thereof is turned off, a spike-like surge current is produced and thus unnecessary loss and noise are generated. However, by providing a dead time, the turn-off can be delayed until the voltage across the both ends becomes 0 V. Consequently, the unnecessary loss and the noise can be suppressed. As a matter of course, the same operation is similarly applicable to the case of a reverse switching process.
The detailed description of each operation mode of the driving signals generated by the driving unit 13 will be omitted (see Patent Document 2). However, the characteristics of the circuit configuration shown in FIG. 9 is that the voltage produced by the first semiconductor switching element 3 and the second semiconductor switching element 4 is equal to the DC power supply voltage VDC, that is, 240√{square root over (2)}=339 V, even in Europe where the highest voltage 240 V is used at general home. Consequently, even though an emergency situation such as lightning surge or abrupt voltage drop is taken into consideration, the first semiconductor switching element 3 and the second semiconductor switching element 4 can be used as a device which has a resistance to a 600 V or so (for example, see Patent Document 2).
Next, FIG. 10 shows a resonant property of this kind in an inverter power supply circuit (where an inductance L and a capacitor C constitute the resonant circuit). FIG. 10 is a diagram illustrating a property of current and a working frequency at the time of applying a predetermined voltage to the inverter resonant circuit, and a frequency f0 is a resonant frequency. During the practical inverter operation, a curved line property I1 (solid line) of the current and frequency is used in the frequency range from f1 to f2 which is higher than the frequency f0.
That is, when the resonant frequency is f0, the current I1 has the maximum, and the current I1 reduces as the frequency range increases from F1 to F3. That is because current which flows in the second coil winding of the leakage transformer increases since the current I1 approaches the resonant frequency at the time when the current I1 approaches the low frequency in the frequency range from f1 to f3. Conversely, since the current I1 becomes more distant from the resonant frequency at the time when the current I1 approaches the high frequency, the current of the second coil winding of the leakage transformer decreases. The inverter power supply for driving the magnetron which is a nonlinear load obtains a desired output by varying the frequency. For example, it is possible to obtain a continuous output, which is not impossible to obtain in an LC power supply, in the vicinity of f3, f2, and f1 in the case of the power output of 200 W, 600 W, and 1200 W, respectively.
In addition, the alternating current commercial power supply is used. Accordingly, when high voltage is not applied to the vicinity of power supply phases 0° and 180°, the inverter operating frequency is configured to the vicinity of f1, where resonant current increases, in the phases depending on a magnetron property in which a high frequency is not oscillated. In this manner, it is possible to increase a conduction angle in which electrical waves are transmitted by raising a boosting ratio of the applied voltage of the magnetron to the voltage of the commercial power supply. As a result, it is possible to embody a current waveform in which the fundamental wave components are numerous and the harmonics components is small, by changing the inverter operating frequency in every power supply phase.
Sequentially, FIG. 11 is a diagram illustrating a property of a change in a temperature of an applied voltage, that is, an oscillation threshold ebm required for the magnetron to irradiate a microwave. The horizontal axis represents anode current Ia that flows after the magnetron oscillates and the vertical axis represents the applied voltage between the anode and cathode of the magnetron. The magnetron is biased to the negative voltage. The applied voltage of about −4 KV is oscillated and the anode current starts to flow and the microwave is irradiated from an antenna. The oscillation threshold ebm of the magnetron is temperature-dependent and is likely to descend, as a temperature is higher.
That is because a magnet is used at the time of the spiral movement of electrons and the magnetism decreases due to the high temperature from 1900 K to 2100 K caused by electronic collision at the time of being oscillated in a cathode unit. To avoid the above phenomenon, it is necessary for the magnetron to be configured as a water-cooling type, such that the change in the temperature becomes very small. However, it is difficult for the general home microwave oven to be configured as the water-cooling type due to an installation condition and cost, and thus the most home microwave oven is configured as a forced air-cooling type. Accordingly, when the temperature increases in the successive movement manner, the oscillation threshold ebm decreases until −3 KV. The solid line shown in the drawing indicates the state of the room temperature and the dashed line indicates the property at the time of the increases in temperature. In this manner, the feedback control of tracking the change in the oscillation threshold ebm caused by the change in the magnetron is important. Above all, the harmonics performance of the power supply depends on how the shape of the frequency modulation waveform is well varied in the initial state of the room temperature, such that the harmonic component is prevented from occurring.
Patent Document 1: JP-A-2004-006384
Patent Document 2: JP-A-2000-058252